Adaptive synchronous switching in a resonant converter

ABSTRACT

An embodiment of a resonant converter includes having resonant circuitry having inductive and capacitive elements configured to create electrical resonance when an input voltage is applied and a synchronous rectifier coupled between at least a portion of the resonant circuitry and an output of the resonant converter. The synchronous rectifier includes a diode, and an electrical switch. Control circuitry is configured to operate the electrical switch such that the electrical switch is turned on when there is substantially no voltage across the diode and current flow in the diode is positive in a direction from anode to cathode.

BACKGROUND

Power electronics are widely used in a variety of applications. Poweradapters having power electronic circuits are commonly used to convertthe form of electrical energy, for example, from AC to DC, from onevoltage level to another, or in some other way. Such devices can operateover a wide range of power levels, from milliwatts in mobile devices tohundreds of megawatts in a high voltage power transmission system.Despite the progress made in power electronics conversion systems, thereis a need in the technology for advanced systems architecture andmethods of operating the same to achieve high efficiencies and improveon size, weight, and complexity of the power electronic devices and itsapplications.

SUMMARY OF THE INVENTION

The present invention relates generally to power electronic converters.More specifically, the present invention relates to resonant converterand adaptive control circuitry. Embodiments may utilize techniquesincluding (1) synchronous switching on resonant circuit primaryswitches, (2) synchronous switching on output synchronous rectifierdrive circuits, (3) operating the resonant converter in “burst mode” tomaintain zero-voltage switching under light to heavy load conditions,and/or (4) active voltage clamping to minimize unnecessary energyclamping that can lead to added component dissipation and reduced powerconverter efficiency.

An embodiment of a resonant converter, according to the disclosure,includes resonant circuitry having inductive and capacitive elementsconfigured to create electrical resonance when an input voltage isapplied a synchronous rectifier coupled between at least a portion ofthe resonant circuitry and an output of the resonant converter. Thesynchronous rectifier includes a diode and an electrical switch. Controlcircuitry is configured to operate the electrical switch such that theelectrical switch is turned on when there is substantially no voltageacross the diode and current flow in the diode is positive in adirection from anode to cathode.

An embodiment of a method of providing electrical power conversion,according to the disclosure, includes providing a resonant converterwith resonant circuitry having inductive and capacitive elements tocreate electrical resonance when an input voltage is applied to theresonant circuitry. The method further includes rectifying an outputvoltage of the resonant converter using a synchronous rectifier coupledbetween at least a portion of the resonant circuitry and an output ofthe resonant converter. The synchronous rectifier includes a diode andan electrical switch. The method also includes operating the electricalswitch such that the electrical switch is turned on when there issubstantially no voltage across the diode and current flow in the diodeis positive in a direction from anode to cathode.

Another embodiment of a resonant converter includes resonant circuitryhaving inductive and capacitive elements configured to create electricalresonance when an input voltage is applied, and a first synchronousrectifier and a second synchronous rectifier. Each of the firstsynchronous rectifier and the second synchronous rectifier includes adiode and an electrical switch in parallel with the diode. Controlcircuitry is configured to operate the first synchronous rectifier andthe second synchronous rectifier such that, for each of the firstsynchronous rectifier and the second synchronous rectifier, theelectrical switch is turned on when current flow in the diode ispositive in a direction from anode to cathode.

Numerous benefits are achieved by way of the present invention overconventional techniques. Methods provided herein enable an AC-DCconverter to operate efficiently while maintaining a desired outputpower level from light to heavy loads. By attaining high efficiency,power system thermal requirement is reduced and power density issignificantly increased. Moreover, the disclosed techniques can assistin preserving the integrity of the switching elements when operating athigh voltages and/or frequencies. Disclosed techniques can apply to bothisolated and non-isolated resonant converters. These and otherembodiments of the invention, along with many of its advantages andfeatures, are described in more detail in conjunction with the textbelow and attached figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram illustrating a non-isolated resonantconverter, according to one embodiment;

FIG. 2 is a drawing of waveforms for V_(S), I_(S1), and Drive 1 of thenon-isolated resonant converter of FIG. 1;

FIGS. 3A and 3B are schematic diagrams illustrating types of feedbackthat can be used to inform the control circuitry;

FIGS. 4 and 5 show waveforms of V_(S) and Drive 1, illustrating burstmode according to one embodiment;

FIG. 5 is a flow diagram illustrating the functionality of anadjustable-output electrical adapter, according to another embodiment;

FIG. 6 shows waveforms illustrating how Drive 1 reduces T_(ON) to reduceoutput power, and how this can result in insufficient current to driveV_(S) back to zero;

FIG. 7 is a schematic diagram illustrating an isolated resonantconverter, according to one embodiment;

FIG. 8 is a simplified illustration of a transformer that can provideleakage inductance, according to one embodiment;

FIGS. 9-12 are schematic diagrams illustrating various embodiments of anoutput stage of an isolated resonant converter;

FIGS. 13-15 are schematic diagrams illustrating various embodiments ofcircuitry for driving synchronous rectifiers, such as those shown inprevious figures; and

FIGS. 16 and 17 are flow diagrams illustrating embodiments of methods ofproviding electrical power conversion.

FIG. 18 is a schematic diagram showing an example controller circuit forproviding S₁ control, according to one embodiment.

FIGS. 19-21 are schematic diagrams showing embodiments of active clampcircuits.

FIG. 22 is a schematic diagram showing active clamp usage in anotherembodiment.

In the appended figures, similar components and/or features may have thesame reference label. Further, various components of the same type maybe distinguished by following the reference label by a dash and a secondlabel that distinguishes among the similar components. If only the firstreference label is used in the specification, the description isapplicable to any one of the similar components having the same firstreference label irrespective of the second reference label.

DETAILED DESCRIPTION

The present invention relates generally to power electronic converters.More specifically, the present invention relates to an adaptive resonantconverter and associated control circuitry. The disclosed embodimentshave the capability to adapt to internal and external changes andoperate under varying line, load, environmental and component parametersand yet preserve very high efficiency ranging from no load to heavyloads. Such a power converter can be utilized, for example, in AC-DCpower converter to power any of a variety of electronic devices such aslaptop computers, USB-powered devices, and the like with very high powerdensities. Techniques detailed herein can apply to both isolated andnon-isolated resonant converters.

Among other things, four methods and techniques are applied in thedisclosed embodiments as follows. 1) A synchronous switching techniqueis used on resonant circuit primary switches. By utilizing zero-voltageswitching, embodiments of the invention can provide for greatly-reducedswitching losses. 2) A synchronous switching technique is also used onoutput synchronous rectifier drive circuits. Control circuitry maymonitor voltage and/or current across certain switches in the primaryand/or secondary circuits to enable such efficient switching. 3)Embodiments may include operating the resonant converter in a “burstmode” to maintain that zero-voltage switching under light to heavy loadconditions. This function is automatically adjusted to compensate forvariations in line and load, also environmental and/or componentparameter changes. Further, the “burst mode” function can be userprogrammable in order to adapt to application and preserve the premiumefficiencies. 4) An active voltage clamping circuit is used to minimizeunnecessary energy clamping that leads to added component dissipationand results in reduced power converter efficiency. This is especiallythe case when isolation transformer in resonant converters is purposelydesigned with high leakage inductance for integration and reducedcomponent counts. The relationship between primary resonant inductance(Lp) and transformer leakage inductance (Llk) is:

Lp=(Llk·Np ²)/Ns ²  (1)

for a 1:1 transformer ratio Lp=Llk. Leakage inductance is selected formaximum power transfer from primary to secondary circuits.

Advanced high density power electronic packaging can be applied toreduce significantly loop inductance of power stages and switchinglosses especially in high frequency resonant converters. Power switchinginterconnections techniques can have benefits in sensing, thermalmanagement, and EMI containment. A fabricated integrated structure bymeans of integrated process flow can be replaced by an assembled one.

Further, part or all techniques disclosed can be applied to a PowerFactor Correction (PFC) or active rectifier circuit feeding a resonantconverter.

Below is description of the above-referenced circuits and techniques.

Synchronous Switching

Control circuitry may monitor voltage and/or current across-circuitswitches to enable synchronous zero-voltage switching. There are manydifferent ways to sense voltage or current. Embodiments are notparticular to a specific method in which voltage or current sensing isaccomplished. For example, current can be measured by hall-effectsensors, precision resistors with or without active circuits forisolation, or current transformers. Sensing primary current alone maynot be a good representative of output resonant current for switchingpurposes. There is a significant phase shift between the primary currentand secondary current that can vary with load, temperature and componentparameter changes.

In some cases one current sensor can be utilized to monitor resonantcurrent and predict current through two or more switches in isolated andnon-isolated resonant circuits. Control may use the desired voltage andcurrent feedbacks to determine best turn-on time of a particular switch.

FIG. 1 is a schematic diagram illustrating a non-isolated resonantconverter 100, according to one embodiment. The embodiment shown in FIG.1 and elsewhere herein are provided as non-limiting examples. One ofordinary skill in the art would recognize many variations,modifications, and alternatives to the components provided herein.

In FIG. 1, the non-isolated resonant converter 100 is configured to takean input voltage V_(IN), which may be an AC, DC, or Rectified AC voltagesource, and provide an output voltage V_(O), which can be greater thanthe input voltage V_(IN). Operation of the non-isolated resonantconverter 100 is determined, in part, by the operation of Drive 1, whichdrives electronic switch S₁, modulating the switch S₁ to meet powerrequirements, thereby making the non-isolated resonant converter 100self-adapting. Inductive elements L_(B) and L_(R) and capacitiveelements C_(P) and C_(R) enable electrical resonance in the non-isolatedresonant converter 100. C_(P) is the lumped circuit capacitance at thenode V, which may include the parasitic capacitance of the semiconductorswitch S₁ and any other electrically connected stray capacitances atthat node such as the L_(B), L_(R) associated capacitances, and thecircuit loading of the L_(R), C_(R) network. Synchronous rectificationat the output is provided by switch S_(R) and diode D_(B), and outputpower is provided across a load resistance R_(L). Additional detailregarding synchronous rectification and Drive 2 are provided hereinbelow.

To enable the non-isolated resonant converter 100 to operate inhigh-power applications at a high switching frequency, specializedcomponents can be utilized. For example, in some embodiments, thetransistors and diodes utilized in the switches can be devices based ona wide bandgap material, such as GaN or silicon carbide (SiC). This canenable the non-isolated resonant converter 100 to operate at highervoltages, higher temperatures, and higher frequencies than solutionsusing traditional silicon-based devices. However, any semiconductordevice can be used.

Specialty magnet materials and geometry may be used to advance the highfrequency operation of the resonant circuits when isolation transformersare used. Advanced material may also be used for other magneticcomponents when non-isolated and isolated topologies are utilized.

The values of the various components utilized in the non-isolatedresonant converter 100 can vary, depending on desired functionality,manufacturing concerns, and/or other factors.

FIG. 2 is a drawing of waveforms for V_(S), I_(S1), and Drive 1 of thenon-isolated resonant converter 100 of FIG. 1, provided to helpillustrate the operation of the non-isolated resonant converter 100. Thewave form of Drive 1 shows how Drive 1 can operate switch S₁ to undergoseveral on/off cycles having switching periods including a time theswitch is turned on, T_(ON), and a time the switch is turned off,T_(OFF).

As shown, when switch S₁ is on, the voltage at the node V_(S) reducesand the current I_(S1) begins to ramp up, reaching a peak current I_(P)when Drive 1 turns the switch S₁ off. When switch S₁ is off, resonancebetween L_(B), C_(P), and equivalent impedance of L_(R) and C_(R)impedance occurs. Current flows into C_(R) where, in some embodiments, avoltage of up to three times the input voltage V_(IN) can be achieved.However, V_(S) then reduces to a lower voltage than V_(IN) due to thereversal of the current I_(S1). The frequency of the resonance will beaccording to the equation:

$\begin{matrix}{{Fr}:=\frac{1}{2 \cdot \pi \cdot \sqrt{L \cdot C}}} & (2)\end{matrix}$

Where L and C are the Thevenin equivalent inductance and capacitance atnode V_(S).

One beneficial feature the non-isolated resonant converter 100 is thatthe voltage waveform created by the switching of S₁ and the resonancecreated between L_(B), L_(R), and C_(P) allows for zero-voltageswitching of S₁. That is, S₁ is switched when the voltage across S₁ isat or near zero volts, which greatly reduces the switching loss in thecapacitance Cp. The voltage across S₁ returns to zero volts when themagnitude of the I_(S1) is great enough, and the voltage is held at zerovolts through the action of the parasitic or intentionally included S₁parallel diode or the subsequent turn on of S₁ at an optimal time T1.

For an optimal turn on of S₁ at T1, the voltage V_(S) can be monitoredeither directly or via a similar representative voltage waveform (e.g.,transformer winding). For example, a zero-voltage-detection circuit canbe utilized such that when V_(S) is approaching or at zero volts, thenext T_(ON) transition for S₁ is started.

The increased efficiency from zero-voltage switching of S₁ node comesfrom the minimization of the energy in capacitance C_(P) which wouldotherwise be dissipated in S₁ according to the formula:

$\begin{matrix}{{EnergyCp}:={\frac{1}{2} \cdot {Cp} \cdot {Vs}^{2}}} & (3)\end{matrix}$

for each switching period.

By quickly turning on S₁ as when the voltage is at or near zero voltsalso reduces and/or eliminates the conduction in the parallel diode(parasitic on intentional) to S₁.

The zero-voltage-detection circuit and optimal turn-on of S₁ at T1 isalso beneficial in that variations in the dynamic or initial values ofthe resonant network (inductance, capacitance or resistance) will changethe shape of the resonant waveform and also change the optimal turn-onstart time T1. However, the zero-voltage-detection circuit can adapt theswitching waveform every switching cycle to help ensure T1 is optimal,largely independent of other circuit variations.

The efficient power transfer from the input to the output (V_(IN) toV_(O)) of a non-isolated resonant converter 100, and similar resonantconverters, is therefore primarily governed by the controlling of S₁ onand off times while maintaining the resonant operation to obtainzero-voltage switching. FIGS. 3A and 3B are simplified illustrationsschematics of how feedback can be used in the control circuit drivingS₁, in various embodiments of a resonant converter.

FIG. 3A illustrates providing feedback in the system around the powerstage with voltage feedback. FIG. 3B, on the other hand, illustratesproviding feedback in the system around the power stage with currentfeedback. It can be noted, however, that other configurations mayutilize a combination of both. For example, embodiments may utilizepower feedback (where P_(O)=V_(O)*I_(O)), in order to control the energyis delivered to the output. As illustrated, the feedback signal in someembodiments may be galvanically isolated (input to output) in anisolated version of the controller scheme, with a signal isolationcircuit (such as an opto-coupler or signal transformer).

Embodiments may potentially utilize a variety of different methods formodulating S₁ to achieve output regulation while maintainingzero-voltage switching. Three such methods include frequency modulation,on-time (T_(ON)) modulation, and pulse density modulation or “burstmode.” This art includes a controlled burst mode to maintainzero-voltage switching under varying internal and external conditions.Details of this feature are described below.

Synchronous Switching Output Stages

Synchronous switching may also be utilized on output stage of a resonantpower converter to further reduce losses and increase efficiency.Synchronous rectification can provide benefits to topologies such asflyback. Example benefits include: 1) Magnetizing current can benegative, hence discontinuous conduction mode is avoided and the outputvoltage is regulated even under no load conditions; 2) zero-voltageswitching can be achieved; and 3) conduction losses of the rectifier aresignificantly reduced specially at low voltage levels.

Below are examples with isolated and non-isolated configurations.

As illustrated in FIG. 1, the non-isolated resonant converter 100 is asimple embodiment of a converter in which isolation is not provided. Asillustrated in FIG. 7 and subsequent figures, however, many variationson the simple design of FIG. 1 can be made, including using circuitrythat provides isolation.

FIG. 7 is a schematic diagram illustrating an isolated resonantconverter 700, according to one embodiment. In this embodiment, atransformer T₁ provides isolation, and may additionally provide voltagechange as well, depending on desired functionality. Double-ended arrowswith dotted lines indicate alternative configurations. Thus, thesecondary winding of the transformer T₁ may be coupled either way,depending, for example, on the desired phase in which the isolatedresonant converter 700 is to be operated, as described in more detailbelow. In some embodiments, the L_(B) inductor of FIG. 1 can becomplemented and/or entirely replaced by the transformer T₁ magnetizinginductance L_(MAG). Additionally, as indicated, the resonant inductorL_(R) can be placed either on the input side (L_(RB)) or the output side(L_(RA)). Alternatively, some embodiments may include both.

The values of the inductive elements can vary, depending on the inputand/or output specifications of the converter. If, for example, theoutput voltage is much lower than the input voltage, the value of theresonant inductance can be reduced by the square of the transformerturns ratio to achieve a much lower value of inductance L_(RA) incomparison to L_(RB). This can facilitate, for example, the use of amuch lower loss air-core inductor instead of one with a magnetic core inposition L_(RB), which would introduce more loss in the inductance.

Inductance on either or both of the input side (L_(RB)) or the outputside (L_(RA)) can also be included in the circuit by the addition ofleakage inductance between the primary and secondary winding. Leakageinductance increases through increasing the physical separation betweenprimary and secondary windings.

FIG. 8 is a simplified illustration of a transformer 800 that canprovide such leakage inductance, according to one embodiment. Accordingto this embodiment, instead of winding primary and secondary windings onthe same core leg (e.g., on top of each other) they can be woundside-by-side or on separate legs to each other, to deliberatelyintroduce a desired amount of leakage inductance. Here, a magnetic core810 is wound with a primary winding 820 on one side, and a secondarywinding 830 on the other side, the magnetic core 810 conducting themagnetic flux 840 between the windings.

This method, when applied to the topology of FIG. 8 (as well as otherembodiments providing isolation), can have at least two significantbenefits. First, the components L_(RA) and/or L_(RB) can be eliminatedas physical components. And second, galvanic isolation can be much moreeasily achieved between primary winding 820 and secondary winding 830.This is the case where the windings are wire-wound or embedded in amultilayer printed circuit board. Further, because the windings are notstacked on each top of other, the number of winding layers is reduced byhalf. This can greatly reduce the cost of a multi-layer printed circuitboard (PCB), in embodiments in which PCB is used for constructingwindings. In embodiments not using PCB, manufacturing costs can still bereduced because windings are relatively easy to wind and the transformerrequires no insulation tape.

In further reference to the isolated resonant converter 700 of FIG. 7,the polarity of the secondary winding 830 of the transistor 800 may bein either direction, which can determine the phase of S₁ on which thepower transfer happens through L_(R) (L_(RA) and/or L_(RB)) and C_(R).

Specialty magnet materials and geometry may be used to advance the highfrequency operation of the isolated resonant circuits. Advanced materialmay also be used for other magnetic components in circuit.

In the isolated resonant converter 700 of FIG. 7, the output diode D_(O)can be moved to a zero voltage reference (i.e., output GND) instead ofthe output positive voltage rail. This means that it can be much easierto include a semiconductor switch rectifier in addition to or inreplacement of a diode rectifier D_(O) because the reference for thedrive signal Drive 2 can be zero volts.

FIG. 9 illustrates an embodiment of an output stage 900 of an isolatedresonant converter. This output stage can, for example, be a variationon the output stage of the isolated resonant converter 700 of FIG. 7.Here, as in FIG. 7, the secondary winding of transformer T₁ may becoupled in either orientation, depending on desired functionality. Inthis output stage, 900 D_(O) is located in the high side (positivevoltage rail) in a configuration similar to the non-isolated resonantconverter. As indicated previously, this configuration is possible, butit can be more difficult to provide a drive waveform for a semiconductorswitch.

Semiconductor switches instead of or in parallel with diode positions(as shown), known as synchronous rectifiers can reduce the conductionlosses of the switch. Such synchronous rectifiers can be included inmany applications where lower resistance is desired. Additional detailregarding Drive 2 is provided herein below.

Note that the position of the capacitor C_(R) in FIGS. 7 and 9 canadditionally or alternatively be across the diode D_(O) position.Because C_(R) is typically much smaller than the output capacitance,C_(R) therefore forms an electrically equivalent circuit when in serieswith the much larger output capacitance. The resonant circuit discussedin previous embodiments therefore also includes the parasiticcapacitance of the diode D_(O).

FIG. 10 illustrates another embodiment of an output stage 1000 of anisolated resonant converter. Again, the secondary winding of transformerT₁ may be coupled in either orientation, depending on desiredfunctionality. This output stage 1000, however, includes a variation inwhich L_(R) (which may also be leakage inductance, as previouslydiscussed) and C_(R) are in series instead of in parallel with thetransformer secondary.

The topology output stage 1000 can be beneficial because the voltage onthe rectifiers D_(O1) and D_(O2) (and/or their semiconductor switchequivalents) is limited to approximately the output voltage plus switchvoltage drop while conducting. In practice, for example, V_(S1) can be3-4 times lower than V_(S) in previous embodiments. This can bebeneficial because lower-voltage-rating diodes and semiconductorswitches can be used. These components typically often have lower onresistance and lower conducting voltage drop, thereby reducing heat andincreasing efficiency. In certain applications, such as high outputcurrent applications, these benefits may justify the increasedcomplexity of output stage 1000.

FIGS. 11 and 12 show additional embodiments of output stages 1100 and1200, respectively. The configurations illustrate electrical equivalentvariations of the output stage 1000 of FIG. 10. FIG. 10 shows how S_(R1)can be coupled in series between the resonant circuitry and the positiverail of the output of the resonant converter, where FIG. 11 shows howS_(R1) can be coupled in series between the resonant circuitry and thenegative rail of the output of the resonant converter. FIG. 12 shows howoutput capacitor C_(O) can be split in two, and the secondary winding oftransformer T₁ can be coupled in-between the two new capacitors C_(O1),C_(O2).

Control for Synchronous Rectifiers

Synchronous switching can be used on output synchronous rectifier drivecircuits. By utilizing zero-voltage switching, embodiments of theinvention can provide for greatly-reduced switching losses. Controlcircuitry may monitor voltage and/or current across certain switches inthe primary and/or secondary circuits to enable such efficientswitching.

For output stage circuits shown in FIGS. 1, 7, and 9, the output stageresonant circuit is parallel, and there is phase difference between thecurrent and voltage waveform. There is also a phase difference betweenthe S₁ waveform and transformer waveform and the output rectifier(D_(O)) position. Therefore, the S₁ controller is unable to determinewhen to turn the synchronous rectifier (S_(R)) on and off.

Assuming that there is a parasitic or intended diode in the S_(R)position, an ideal condition for turning on switch S_(R) occurs when thediode voltage is minimized (conducting) and the current flow in thediode is positive (anode to cathode). The voltage information may not beenough on its own to operate S_(R) because once the S_(R) turns on andvoltage is minimized, it will be difficult to determine when to turnS_(R) off from the low D_(O)/S_(R) voltage.

FIG. 13 is a schematic diagram illustrating a block level solution tothis problem. Here, a current transformer, AND gate, inverter amplifier,and a driver to help ensure that S_(R) is turned on when the voltage isminimized (e.g., at or approaching zero) and the current flow inD_(O)/S_(R) is positive (anode to cathode direction). S_(R) is turnedoff again when the current flow is approximately zero. Here, the primarywinding of the current transformer is coupled to a current of at least aportion of the resonant circuitry (e.g., I_(R), I_(DB), or I_(DO)), theinverter amplifier is coupled to a node of the diode D_(O), and the ANDgate is configured to perform a Boolean AND function using an output ofthe current transformer (e.g., from the secondary winding) and theoutput of the inverter amplifier. This circuit can be used to providethe signal for Drive 2 in FIGS. 1, 7, and 9.

As for the circuits shown in FIGS. 10-12, because the capacitor C_(R) isin series with the Do element, it may only be necessary to know when thecurrent flow is positive in the rectifier (anode to cathode).Accordingly, the circuit in FIG. 14 shows a solution for determining thecontrol drive for S_(R1) and S_(R2) (individually).

Alternatively, the current in C_(R) may be sensed bi-directionally witha single current transformer as the positions S_(R1) and S_(R2) conductin opposite phases either with two anti-phase secondary windings. Anexample of such a circuit is provided in FIG. 15. Here, the current inthe secondary resonant circuit is detected using first and secondsecondary windings of a current transformer, which drive switches S_(R1)and S_(R2) via drivers. When the current is positive, one rectifier isturned on, and when the current is negative, the other is turned on.Alternatively, a single current transformer secondary winding withpositive and negative current detection (not shown) can be utilized.

FIG. 16 is a flow diagram illustrating a method of providing electricalpower conversion, according to one embodiment. The functionality, inwhole or in part, can be provided by hardware and/or software, includingthe circuitry and other components described in relation to FIGS. 1, 7,and 9-12.

At block 1610, a resonant converter is provided with resonant circuitryhaving inductive and capacitive elements to create electrical resonancewhen an input voltage is applied to the resonant circuitry. Values ofinductive and capacitive elements can vary, depending on switchingfrequency, desired functionality, and/or other factors.

At block 1620, voltage-monitoring circuitry is used to determine whenthere is substantially no voltage across an electrical switch coupled tothe resonant circuitry. As illustrated in FIGS. 1-6, the control of aswitch (e.g., switch S₁ of FIG. 1) can be based on the voltage acrossthat switch. Switching efficiency is optimized when the voltage is at ornear zero. Accordingly, at block 1630, the electrical switch is operatedsuch that the electrical switch is turned on when substantially novoltage is detected across the electrical switch.

Optionally, at block 1640, the electrical switch is operated in a modein which the switch undergoes a plurality of on-off cycles over a periodof time before being turned off A “burst mode” can allow the resonantconverter to maintain output power while enabling for zero-voltageswitching.

FIG. 17 is a flow diagram illustrating a method of providing electricalpower conversion, according to another embodiment. Similar to FIG. 16,the functionality shown in FIG. 17, in whole or in part, can be providedby hardware and/or software, including the circuitry and othercomponents described in relation to FIGS. 1, 7, and 9-12.

At block 1710, resonant converter is provided with resonant circuitryhaving inductive and capacitive elements to create electrical resonancewhen an input voltage is applied to the resonant circuitry. Again,values of inductive and capacitive elements can vary, depending onswitching frequency, desired functionality, and/or other factors.

At block 1720, an output voltage of the resonant converter is rectifiedusing a synchronous rectifier comprising a diode and an electricalswitch. Such rectification is provided in the previously-discussedembodiments, for example, by switch S_(R). As discussed above, switchingefficiency for synchronous rectifiers may be timed not only based onvoltage, but on current as well. Thus, at block 1730, the electricalswitch is operated such that the electrical switch is turned on whenthere is substantially no voltage across the diode and the current flowin the diode is positive in a direction from the anode to the cathode.

It should be appreciated that the specific blocks shown in FIGS. 16 and17 illustrate methods of providing electrical power conversion accordingto two specific embodiments. Other embodiments may include alternativeand/or additional functionality. Embodiments may further includefunctionality that is not illustrated in FIGS. 16 and 17. Furthermore,steps may be added, removed, and/or rearranged depending on theparticular applications. One of ordinary skill in the art wouldrecognize many variations, modifications, and alternatives.

It will be understood the examples and embodiments describing“zero-voltage” switching may not operate switches at exactly zerovoltage. Different tolerances of components and materials used in thecircuitry can cause, for example, a zero-voltage detector to vary in itsdetection of zero volts. However, such a detector may detect a voltageof substantially zero (i.e., substantially no voltage), where anyexisting voltage is, within tolerances, considered zero volts forpurposes of which it is used.

It is also understood that the examples and embodiments described hereinare for illustrative purposes only and that various modifications orchanges in light thereof will be suggested to persons skilled in the artand are to be included within the spirit and purview of this applicationand scope of the appended claims.

Controlled Burst Mode

Embodiments may potentially utilize a variety of different methods formodulating S₁ to achieve output regulation while maintainingzero-voltage switching, including a controlled “burst mode.” An overviewof such modulation techniques are described below.

Frequency Modulation.

With frequency modulation, the higher the frequency the lower the valueof I_(P), which means that switching frequency can be used to regulateoutput power. However, frequency regulation typically excludes the useof a zero-voltage detector, which (zero-voltage detecting method) is inconflict with frequency modulation because it can change the T_(OFF)time (duty-cycle) and consequently the switching period (and thereforefrequency).

Maximum T_(ON) Modulation.

In accordance with maximum T_(ON) modulation, S₁ can be modulated suchthat Ton has a maximum on time proportional to 1/V_(1N). That is, thehigher the input voltage V_(IN), the shorter the length of Ton. Thishelps ensure that the maximum power transfer in the circuit isrelatively constant with any variation in V_(IN), as I_(P) is closelyrelated to T_(ON). Although this may be the case for maximum powertransfer (maximum output load) further T_(ON) modulation may benecessary to regulate the output (voltage, current, or power) tolower/light output loads. Additional details regarding optimal T_(ON)modulation are provided herein below.

Controlled Burst Mode.

A maximum time for T_(ON) may be preferentially proportional to1/V_(1N). Here, however, the switch S₁ is driven on and off for burstintervals, rather than continually. In this way the average powertransferred is reduced.

FIGS. 4 and 5 show waveforms of V_(S) and Drive 1, illustrating burstmode according to one embodiment. FIG. 4 illustrates waveforms of asingle series (or “burst”) of on/off cycles for S₁. Output power can bemaintained and/or adjusted by adjusting the frequency of bursts. FIG. 5illustrates an example of how bursts be provided in succession tomaintain a certain output power. Additionally, as indicated above,T_(ON) may be adjusted to maintain a certain output power andzero-voltage switching.

As FIG. 4 illustrates, the on time T_(ON) of each on/off cycle in aburst mode can be progressively longer. These increasing T_(ON) periodsare labeled B1, B2, B3, and B4. By increasing the lengths of T_(ON) asDrive 1 progresses from B1 to B4, the resonant network is able toprogressively establish resonance for each burst without overshoot.Without using such progressive modulation, the initial resonant peaks ofV_(S) following B1, B2 etc. would be much higher and overshoot couldresult, which could be harmful for the switch S₁.

It will be understood that the waveforms of FIGS. 4 and 5 are providedfor illustrative purposes. In practice, various features of theillustrated waveforms, such as number of switching periods in a singleburst period, magnitude of V_(S) and Drive 1, the duty cycle of eachon/off cycle, and the like can vary, depending on the configuration,power requirements, and/or other factors.

Further, the “burst mode” function can be made user programmable inorder to easily adapt to particular application and preserve the premiumefficiencies. It can be beneficial to initiate burst mode when zero-volthas not been achieved in resonant circuits. This is especially the casefor a wide range applications in which the converter operates tominimize power dissipation at light loads to achieve high efficiencies.Without a mechanism to detect zero-volt, switches can be damaged atlight loads when switching at very high frequencies.

In accordance with some embodiments, the switch S₁ can be a GaNtransistor, such as a MOSFET, MESFET, and the like. In such embodiments,the switch S₁ can be modulated at much higher frequencies than similarsilicon-based switches. Higher-frequency switching allows for areduction in size of magnetic and capacitive components, which canreduce the overall size and cost of the power adapter. In someembodiments, for example, the order of magnitude of the switchingfrequencies can be in the megahertz, while bursting frequencies can bein the tens of kilohertz.

T_(ON) Modulation.

As a variation to the burst mode described above, T_(ON) can becontrolled for lower output loads to achieve the required set-point andregulation. (That said, some embodiments may us T_(ON) modulationtogether with other modulation techniques.) Although T_(ON) modulationcan be successful in the output power range of most applications, atlower output loads the T_(ON) time will be small. A smaller T_(ON) timecan result in a smaller current I_(P). And in certain circumstancesthere may not be enough circulating energy in the resonant network forV_(S) to return to zero-volts. FIG. 6 helps illustrate this dilemma, aswell as a solution that can be implemented, according to someembodiments.

FIG. 6 shows waveforms illustrating how, using T_(ON) modulation, Drive1 reduces T_(ON) to reduce output power, and how this can result ininsufficient current I_(S1) (not shown) to drive V_(S) back to zero.Here, a zero-voltage detection circuit can be used to recognize whenzero-voltage switching fails to occur and help the circuit preventefficiency loss and potential damage to the switch S₁ that could resultfrom switching when V_(S) is not at or near zero.

In the illustrated example, the zero voltage detection signals can bemonitored to determine when there is failure of V_(S) to return to zero.If, for example, the zero-volt-detection signal is not received for anumber of switching cycles, Drive 1 is disabled for a period of time sothat when Drive 1 is enabled again, the circuit requires moreinstantaneous power to regulate the average power to the required level.The increase in power requirements enables Drive 1 to have longer T_(ON)times, which allow for zero-voltage switching again for the next numberof cycles.

In other words, a burst mode can be initiated cyclically whenzero-voltage switching of V_(S) is detected to have failed.

Methods for detecting zero-voltage enable adaptive control to enhanceburst mode operation and preserve converter efficiency at allloads—especially light loads—and prevent potential switch damage.

FIG. 18 shows an example controller circuit for providing S₁ control,according to one embodiment. A 555 timer is used in a stable mode togenerate circuit resonant frequency. R_(A), R_(B) and C values determinethe frequency. The 555 timer's output frequency and zero-volt detectsignals are fed to an OR gate which then it would trigger the timeroutput. A zero-volt detect signal is necessary to set the timer outputhigh. Output of the 555 timer is fed to Driver 1. Output voltage is alsosensed and fed to a comparator for voltage regulation. The 555 timer istherefore reset in case the output voltage is higher than reference orother protection signals are activated low.

Controlled Active Clamping

A controlled active clamping technique can be used to hold peak resonantvoltage at pre-determined levels in order to force zero-volt switchingand prevent possible switch damage due to stress voltage, as well aseliminate unnecessary clipping that leads to excess losses and converterinefficiency. In an isolated converter, when transformer peak resetvoltage is significantly larger than the input voltage, a clamp circuitis activated at predetermined peak voltage. Modulation can reduce excessloss in the clamp circuit under varying load conditions. The modulationof peak voltage allows for efficient power transfer and controllableoutput voltage regulation.

Typically snubbing and clamping circuits, such asresistor-capacitor-diode (RCD) circuits, are used on switches to limitvoltage spikes to reduce component stress. This leads to extra circuitdissipation, and thus power savings can be realized. In such circuits, avoltage spike is caused by the energy stored in the transformer'sleakage inductance of an isolated resonant circuit, when the switchturns off and abruptly halts current flow in the primary winding. Thefirst step to reducing both the voltage spike and the loss in the clampis to design a transformer with minimal leakage inductance, which maynot be ideal for a resonant converter. The resonance between thisinductance and the parasitic capacitance of the switch produce largevoltage stress as well as losses, therefore decreasing converterefficiency. The clamp resistance can be increased to further reduce theloss, but doing so also increases the magnitude of the voltage spike.During the reset portion of the switching cycle, the reflected outputvoltage is impressed across the clamp resistor leading to extra loss.Using a higher voltage switch provides more margin for the voltage spikeand allows for a much larger resistor. However, an increased voltagerating results in higher on-resistance which leads to lower efficiencyat high loads. When a controller is operating in burst mode, the clampcircuit discharges between ON states. If the clamp capacitor is toolarge, excess energy is stored and dissipated during the OFF state. Insome situations, the clamp capacitor may not fully discharge before thenext ON state begins.

Embodiments can utilize an active clamping technique rather than an RCDclamp circuit. A non-dissipative LC plus clamp switch circuit, forexample, can to force the transformer leakage inductance energy tooscillate on input as reactive power and/or transfer the energy to loadas real active power. In either case, the energy is not dissipated in aresistor and the losses are decreased. Benefits to an active clampcircuit include the ability to transfer energy under wide line and loadvariations. The technique is suitable for resonant circuits includingPower-Factor Correction (PFC) circuits. Transformer reset isaccomplished with an active clamp circuit consisting of switch and acapacitor working with transformer leakage inductance. Active clampcircuit works as a controllable current source, so as to regulate poweraccording to load variations.

This arrangement offers many benefits. For example, the duty cycle cango higher than 50%, resulting in higher turns ratio, lower primarycurrents and secondary voltages, and smaller output inductor. Also, thevoltage stress on the primary switch remains relatively constant overthe full input voltage range, leading to better overall efficiency. Inaddition, zero-volt switching is possible with this approach, which canlead to further size reduction by increasing the switching frequency.

FIGS. 19-21 are schematics of examples active clamp circuit that can beutilized in embodiments for isolated converters. Values for the variouscomponents involved, alterations to the architecture, and othervariations can vary, depending on desired functionality and will beunderstood by a person of ordinary skill in the art. As shown, thecircuits of FIGS. 19 and 20 utilize a comparator and driver to determinewhen the active clamp switches on, which, as described above, can occurat any of a variety of desired voltages, depending on application anddesired functionality (e.g., 500 V, 800 V, etc.). The circuit of FIG.20, on the other hand, illustrates how clamp switch can be fed by awinding of the transformer between nodes 2 and 3. Thus, the turn-onvoltage for the clamp switch can be determined by number of windingsbetween nodes 2 and 3. The circuit in FIG. 21 therefore illustrates howactive clamping can be done with passive components.

FIG. 22 illustrates a technique applied to a PFC circuit similar to thenon-isolated circuit of FIG. 1. In a low-power, low-current application,using a small magnetizing inductance to achieve zero-volt switching canbe more appropriate. Resonance between the leakage inductance and theclamp capacitance takes place when the transformer is reset. Magnetizinginductance is designed together with the switching frequency in order toprovide zero-volt switching at high input voltage and keep the size andlosses in the transformer small.

When transformer peak reset voltage is significantly larger than theinput voltage, the clamp circuit can be activated at a predeterminedpeak voltage. Modulation can reduce excess loss in the clamp circuitunder varying load conditions. Modulation of peak voltage allows forefficient power transfer and controllable output voltage regulation.Having the ability to control the peak voltage level clamping alsoallows for zero-volt switching of S₁ and also the clamp switch. Clampcircuit and burst mode control force zero-volt switching under varyingload conditions, especially light loads.

The voltage across S₁ can be sensed using techniques similar to themethods shown in FIGS. 19-2120. Other methods can also be applied forsensing voltage. The sensed signal is compared to a reference voltage bycomparator in the clamping circuit. At a predetermined OFF state voltage(i.e., a threshold voltage), clamping switch is turned on, and excessresonant energy is put back to the V_(BUS). The adaptive nature of thecircuit makes it possible to compensate for load and environmentalvariations to achieve higher efficiencies.

Having described various embodiments of the invention, it will beunderstood that the examples and embodiments described herein are forillustrative purposes only and that various modifications or changes inlight thereof will be suggested to persons skilled in the art and are tobe included within the spirit and purview of this application and scopeof the appended claims.

1-18. (canceled)
 19. A non-isolated resonant converter comprising:resonant circuitry having inductive and capacitive elements configuredto create electrical resonance when an input voltage is applied; a firstelectrical switch coupled to the resonant circuitry such that the firstelectrical switch conducts a current of the resonant circuitry;voltage-monitoring circuitry coupled to the resonant circuitry andconfigured to determine when there is substantially no voltage acrossthe first electrical switch; and control circuitry configured to:receive an input from the voltage-monitoring circuitry, and operate thefirst electrical switch; wherein the control circuitry is configured toturn the first electrical switch on when substantially no voltage isdetected across the first electrical switch.
 20. The non-isolatedresonant converter of claim 19 wherein the first electrical switchcomprises a GaN transistor.
 21. The non-isolated resonant converter ofclaim 19 wherein the control circuitry is further configured to operatethe first electrical switch in a mode in which the first electricalswitch undergoes a plurality of on/off cycles over a period of timebefore being turned off.
 22. The non-isolated resonant converter ofclaim 21 wherein the control circuitry is further configured to operatethe first electrical switch such that, for each on/off cycle of theplurality of on/off cycles, a time the first electrical switch is turnedon is progressively longer with each successive on/off cycle.
 23. Thenon-isolated resonant converter of claim 21 wherein the controlcircuitry is further configured to periodically operate the mode tomaintain a certain output power.
 24. The non-isolated resonant converterof claim 19 wherein the control circuitry comprises modulationcircuitry.
 25. The non-isolated resonant converter of claim 24 whereinthe modulation circuitry is programmable.
 26. The non-isolated resonantconverter of claim 19 wherein the control circuitry is furtherconfigured to receive a voltage feedback from an output of thenon-isolated resonant converter.
 27. The non-isolated resonant converterof claim 19 wherein the control circuitry is further configured toreceive a current feedback from an output of the non-isolated resonantconverter.
 28. The non-isolated resonant converter of claim 19 furthercomprising a synchronous rectifier between the resonant circuitry and anoutput of the non-isolated resonant converter, wherein the synchronousrectifier comprises: a diode; a second electrical switch in parallelwith the diode; and switching circuitry configured to operate the secondelectrical switch such that the second electrical switch is turned onwhen substantially no voltage across the diode is detected and currentflow in the diode is positive in a direction from anode to cathode. 29.The non-isolated resonant converter of claim 28 wherein the switchingcircuitry is further configured to operate the second electrical switchsuch that the second electrical switch is turned off when the currentflow is substantially zero.
 30. A method of providing electrical powerconversion, the method comprising: providing a resonant converter withresonant circuitry having inductive and capacitive elements to createelectrical resonance when an input voltage is applied to the resonantcircuitry; using voltage-monitoring circuitry to determine when there issubstantially no voltage across an electrical switch coupled to theresonant circuitry; and operating the electrical switch such that theelectrical switch is turned on when substantially no voltage is detectedacross the electrical switch.
 31. The method of claim 30 furthercomprising operating the electrical switch in a mode in which theelectrical switch undergoes a plurality of on/off cycles over a periodof time before being turned off.
 32. The method of claim 31 furthercomprising operating the electrical switch such that, for each on/offcycle of the plurality of on/off cycles, a time the electrical switch isturned on is progressively longer with each successive on/off cycle. 33.The method of claim 30 further comprising operating a synchronousrectifier coupled to the resonant circuitry and an output of theresonant converter, the synchronous rectifier having a second electricalswitch coupled in parallel to a diode, wherein operating the synchronousrectifier comprises operating the second electrical switch such that thesecond electrical switch is turned on when there substantially novoltage across the diode is detected and current flow in the diode ispositive in a direction from anode to cathode.
 34. The method of claim32 further comprising operating the second electrical switch such thatthe second electrical switch is turned off when the current flow isdetected to be zero.
 35. A resonant converter comprising: an input stageconfigured to receive an input voltage and comprising a first electricalswitch coupled in series with a primary winding of a transformer; anoutput stage configured to provide an output voltage and comprising acapacitive element coupled to secondary winding of the transformer suchthat electrical resonance can occur when the input voltage is applied;voltage-monitoring circuitry coupled to the first electrical switch andconfigured to determine when there is substantially no voltage acrossthe first electrical switch; and control circuitry configured to:receive an input from the voltage-monitoring circuitry, and operate thefirst electrical switch; wherein the control circuitry is configured toturn the first electrical switch on when substantially no voltage isdetected across the first electrical switch.
 36. The resonant converterof claim 35 wherein either or both the input stage or the output stageincludes an inductive element configured to provide the electricalresonance together with the capacitive element.
 37. The resonantconverter of claim 35 wherein the control circuitry is furtherconfigured to operate the first electrical switch in a mode in which theelectrical switch undergoes a plurality of on/off cycles over a periodof time before being turned off.
 38. The resonant converter of claim 37wherein the control circuitry is further configured to operate the firstelectrical switch such that, for each on/off cycle of the plurality ofon/off cycles, a time the first electrical switch is turned on isprogressively longer with each successive on/off cycle.
 39. The resonantconverter of claim 35 wherein the output stage further comprises asynchronous rectifier coupled to a node of an output of the resonantconverter, wherein the synchronous rectifier comprises: a diode; asecond electrical switch in parallel with the diode; and switchingcircuitry configured to operate the second electrical switch such thatthe second electrical switch is turned on when there is substantially novoltage across the diode and current flow in the diode is positive in adirection from anode to cathode.
 40. The resonant converter of claim 35wherein: the output stage further comprises a first synchronousrectifier and a second synchronous rectifier, wherein each of the firstsynchronous rectifier and the second synchronous rectifier comprise: adiode; and an electrical switch in parallel with the diode; and theresonant converter further includes switching circuitry configured tooperate the electrical switch of each of the first synchronous rectifierand the second synchronous rectifier such that, for each of the firstsynchronous rectifier and the second synchronous rectifier theelectrical switch is turned on when current flow in the diode ispositive in a direction from anode to cathode.
 41. The resonantconverter of claim 35 further comprising clamping circuitry to control avoltage across the first electrical switch.
 42. The resonant converterof claim 41 wherein the clamping circuitry comprises active clampingcircuitry in which a clamping switch is turned on when a voltage acrossthe first electrical switch reaches a threshold voltage.
 43. Theresonant converter of claim 41 wherein the clamping circuitry comprises:a clamp capacitor; and an electrical clamp switch coupled in series withthe clamp capacitor; a sensor configured to measure a voltage across thefirst electrical switch; a comparator circuit coupled with an output ofthe sensor and configured to compare the voltage across the firstelectrical switch with a reference voltage; and a driver coupled to anoutput of the comparator circuit and configured to turn on theelectrical clamp switch.
 44. The resonant converter of claim 35 furtherwherein the control circuitry comprises modulation circuitry.